Frequency modulation transmitter



Sept. 27, 1949. F. w. FRINK 2,483,271

FREQUENCY MODULATION TRANSMITTER Filed Nov. 30, 1946 6 Sheets-Sheet 1 COMPENSATING NETWORK FIG. 1 (A) INVENTOR FREDERICK W. FRINK ATTORNEY 6 Sheets-Sheet 2 RF. OUTPUT B 7 0 2 w [4 s a a, o 8 RM v 8 3 II I O m E m a o (WP mF.Wm WW m a R E. ma m Sept. 27, '1 949. F. w. FRINK FREQUENCY MODULATION TRANSMITTER Filed Nov. so, 1946 FIG- 1 (B) Sept. 27, 1949. F. w. FRINK 2,483,271

FREQUENCY MODULATION TRANSMITTER Filed Nov. so, 1946 a Sheets-Sheet s FREDERICK W. FRINK BY zdbwm ATTORNEY Sept. 27, 1949." F. w. FRINK 2,483,271

- FREQUENCY MODULATION TRANSMITTER Filed Nov. 30, 1946 6 Shee'ts-Sheet 5 In v v v w T gae as L so IIIH +I 'IIH M in g W 13% l 5 3H my IHI NN/" Hill FIG- IO I ATTORNEY Sept. 27, 1949.

Filed Nov. 30, 1946 VOLTAGE ACROSS RESISTOR 49 FIG.

FREQUENCY MODULATION TRANSMITTER VOLTAGE ACROSS RESISTOR 3l2 FIG, I2

F. W. FRINK VOLTAGE ACROSS RESISTOR 332 6 Sheets-Sheet 6 FIG. l3

TNVENTOR FREDERICK w. FRINK ATTORNEY Patented Sept. 27, 1949 UNITED STATES PATsNT OFFICE 2,483,271 FREQUENCY MODULATION TRAN SMITTER Frederick W. F rink, East Orange, N. J Application November 30, 1946, Serial No. 713,219

13 Claims. (Cl. 33224) This invention pertains to a system and apparatus for the transmission .of frequency-modulated radio signals. More particularly it pertains to radio transmitters of the type commonly referred to as amplitude to phase to frequencymodulated.

In transmitters of the type just referred to, the original modulation is of the amplitude type. This original modulation is caused to vary the phase of the carrier wave, such phase modulation giving rise to frequency-modulation, as is well known in the art.

A typical prior art system of this type passes the original audio-frequency through a corrective network having such characteristics that the output thereof is inversely proportional to the frequency of the audio-frequency voltage. The output of this network is supplied to a balanced modulator, Where it modulates the radio-frequency current also supplied to this modulator, this latter being usually derived from a crystal oscillator. The modulated output of the balanced modulator is combined with unmodulated current from an identical source, after one or the other current has been shifted 90 degrees in phase, thus yielding phase modulation.

The system just described is capable of produc ing a base deviation approaching :90 degrees, but in practice it is impossible to extend the phase deviation to such a value, since extremely serious distortion would thereby occur. The fact that the phase angle is not directly proportional to the amplitude .of the audio-frequency modulating signal is responsible for this distortion. In devices of the types in actual use, the extent of the deviation has been limited to values arbitrarily postulated as being the permissible maxima which can be used, while at the same time maintaining the distortion below a level which is considered permissible for a particular type of transmission. For example, high-fidelity broadcast transmitters usually limit the deviation to a maximum of 111.5 degrees, which is equivalent to 10.2 radian. Intelligible speech transmission may usually be obtained with deviations as great as :25 degrees, equivalent to $0.44 radian.

In frequency-modulation transmitters it is necessary to secure an ultimate deviation much greater than that corersponding to the values just given. In broadcast transmitters, there is usually desired at the ultimate output a 75-kilocycle deviation. In orderto secure this, the out-'- put of the modulation system passes through an extensive cascaded chain of frequency-multipliers. For example, with a modulating frequency of 30 cycles per second, there initially results a 'requency deviation of only 6 cycles per second, if the deviation be kept within the limits imposed by permissible distortion, as just stated, whence the need for extensive increase in deviation.

The employment of frequency-multipliers allows a crystal oscillator of 200 kc., for example, to be used as the primary frequency source, such an element being readily available. In order to secure the required output frequency, which latter is usually in the vicinity of 100 megacycles, the frequency-multipliers are employed. At the same time, the frequency multiplication thus obtained is still insufiicient to yield the desired kc. ultimate deviation, as the ultimate resting frequency simultaneously produced by the multiplication needed for 75 kc. deviation would be far above the order of megacycles.

As one commonly used solution of this difficulty, the modulated signal is frequency-multiplied to a convenient value, then heterodyned to a lower value, which heterodyning does not reduce the absolute value of the deviation. Further frequency multiplication then simuitaneously yields the desired ultimate resting frequency and the desired ultimate frequency deviation. However, such a solution poses several problems and introduces additional complications and difficulties. The need for extremely accurate control of more than one oscillator is a serious problem, and the various artifices which have been used to correct or compensate for oscillator frequency drift are characterized by the introduction of still further complexity in the apparatus, as typified by-the well known dual-channel system.

The present invention overcomes the difiiculties just recited by providing a phase modulator which produces an initial deviation so great that the number of frequency multipliers needed will be much less than the number ordinarily needed When a phase modulator of conventional type is used. Therefore, it to employ heterodyning, or dual-channel operation, while at the same time the distortion accompanying this increased phase deviation is kept at an extremely low value.

As one example, by the employment of apparatus according to this invention, initial phase deviation may extend to at least 1-11 radians, without introducing distortion above the'max'imum allowable, and theoretically this deviation may be extended to an even greater extent, if desired, although the necessary apparatus for efiecting a greater range becomes relatively extensive. However, 11 radians deviation allows will no longer be necessary .are completely avoided.

the usually desired respective values, as above stated, of ultimate resting frequency and ultimate deviation to be obtained without recourse to heterodyning operations, and with a much lower degree of frequency multiplication than that usually required, which latter is in the neighborhood of '7 or 8 thousandfold.

One object of this invention is to provide a frequency modulation transmitter which will require the. employment of a much smaller amount of frequency multiplication than has hitherto been found necessary in order to produce the requisite degree of ultimate phase deviation.

Another object of this invention .is to provide a frequency modulation transmitting system in which an initial phase deviation :many fold greater than that hitherto employed is .used, while distortion is yet maintained below the maximum permissible in practice.

Yet another object of this invention is to pro- .videa frequency-modulatedtransmitter in which :complexity .andconsequent cost are reducedby reason-of a substantial lessening of the degree of frequency multiplication required, while ;at the same-timethe standards required in practicewith ,respect to distortion and ultimate resting frequency and phase deviation are met.

A further ,purpose of this invention is to provide a frequency-modulation transmitter in which heterodyning operations and the consequent employment of more than one :oscillator -A un further object -of this invention is to provide a frequency-modulation transmitter in --which the audio-frequencycurrentsare somodi- .fied, and modulate the radio-frequency output of ;a single oscillator in such fashion, that modulation of the order of at least twenty times that permissible Joy hitherto used methods, .may be obtained without substantial distortion.

An additional purpose of a variant form 'ofrthis invention is to provide a transmitter of the type iust identified, in which the audio-frequency .currents are still further modified, so that the degree of ultimate modulation secured may beincreased to an even greater value.

Qther objects and features of the invention will be apparent from the following detailed ;description with reference to the accompanying d awin n which:

Figure 1 is a schematic diagram-of one embodiment of my invention, comprising Sheets hand B,

Figures ,2, 3,4, 5,,and 6 are graphs showing :the voltage variations occurring in certain .circuits of the embodiment shown ,in Figure '1,

Figures '7, 8 and 9 are vector diagrams showing the phase conditions existing at certain times in one of the circuits of Figure 1,

Figure 10 is a schematic diagram of a portion .of another embodiment .of this invention, incorporating elements for producing additional phase deviation, and 7 Figures 11, 12 and 13 are graphs showing the variations of potential which occur at certain points in the embodiment shown in Figure 10.

Referring now to Figural, the audio-frequency signal voltage to be transmitted is passed through a compensating network similar to those whioh have already been used in the prior art irequency-modulation transmitters that employ phase modulation circuits. This compensating network consists of resistor 1:0 and capacitor 44, and the circuit constants are so chosen that the ratio of output voltage to input voltage is in .Iidentical secondary windings, it'll-and :15, each of which is tapped substantially at its midpoint. The midpoint of secondary winding I4 is con- :nected to the positive end of battery it, which hasits negative end, grounded, and the midpoint of secondary winding 15 is connected to the anegativeiendbof battery H, which latter has its ,positivaendgrounded. Each of these batteries has a potential of about 40 volts, for the particular-embodiment here shown. It is to be understood that the showing of batteries is purely exemplary, and that any other suitable potential source may be substituted therefor. This :mode of connection allows a direct-current supply of .energy to be impressed upon voltage divider l8,

.as well as upon other voltage dividers hereinafter described, from a single potential source. 'Atthe ,same atime, this particular connection allows such potential source .to be grounded, or to have a relatively high capacity to ground, this last requirement being-one which {it would be difficult rtorneet, if individual batteries wereemployed for each voltage divider.

In the following description, it is assumed, for thesake of simplicity, that the form of the audio-frequency voltage is that of a sine wave .at-the maximum energy level.

Avoltage divider l8,-whi ch consists of one ,or more resistors having ;a total of four taps, is tconnectedirom the'upper-end of secondary winding I4 to the upper end of secondary winding 15. Current from by-passed :batteries 56 and H ;-fiows through this voltage divider, producing an overall D.-C. drop :of about volts, and the ,tap marked 0 is at -D.-C. ground potential. The ll-1C. potentials of the other .taps on this voltage divider relative to ground .are indicated .by [the respective numerals on the diagram. Voltage divider .19, which is similar to H3, con- ;nects the lower end of secondary winding M to the lower end of secondary winding 1 5.

The transformer secondary windings are wound in .such a :direction that the .A.-C. potential of :the :upper end of secondary winding 14 relative to ground is equal, in magnitude .and direction, :to the A.-C. potential of the .upper end of secondary winding 15 relative to ground. There is, therefore, neveranyArC. voltage applied across voltage divider :IS :byztransformer I3. Similarly, there is .never any A.C. voltage applied across voltage divider l9. All points on voltage divider :18 are, therefore, at the same A.- C. potential relative to ground, and a similar statement applies to IS.

The various taps on voltage dividers I8 and I9 are connected, through resistors '2! to 128 inclusive, to diodes :3I to 38 inclusive, and these diodes are connected to bus M, which latter is initurn connected through resistor 49 to ground. To minimize the number of separate tubes renuired, four twin diodes may be used. Each twin diode consistsoftwo independent diodes enclosed in the :same envelope. However, wholly independent diodes may alternatively be used.

Let us :suppose that the instantaneous voltage e existing across one half the respective secondarywinding I4 or |5 has just passed through zero, and. is now increasing in such a direction that the upper-end of I5 is positive with respect to the midpoint. Diode 3| immediately draws current through resistors, 2| and 49, since the plate ofthis diode is at ground potential only when es is zero; It is assumed for illustrative purposes that the resistance of voltage divider l8 and the internal resistances of the various diodes are. negligible compared with the resistances of resistors 2| to28, and d9, inclusive. It is also assumed that the values of resistors 2| and 49 have been so chosen that when es has risen to an instantaneous value of 10 volts the voltage drop e0 across 49 will be 4 volts. If es continues to increase beyond 10 volts, diode 32 starts to draw current through resistors 22 and 49, because the transformer voltage 65 is then greater than the algebraic sum of co (4 Volts) and the D.-C. cathode bias voltage of diode 32 (14 volts), which have opposing polarities so far as diode 32 is concerned;

. The resistance value of resistor 22 is so chosen that thecurrent, through 22 builds up much faster than the current through 2|, and by the time as has reached a value of volts the current through 22 and 32 has reached a magnitude great .enough to reduce the resultant current through 49 to zero; thus reducing voltage co to zero.

The'method of determining correct resistance values for all of the resistors will be later explained, but at presentit is merely assumed that when es increases above 20 volts, diode 33 starts drawing current, and this current builds up fast enough so that by the time es arrives at a value of volts the voltage e0 across 49 will again have risen to a value of 4 volts. Furthermore, when es increases above 30 volts; diode 34 starts drawing current, and by the time e5 reaches a value of volts this current has reached a high enough value so that the voltage e0 has again been reduced to zero.

Reference is now additionally made to Figure 2, where the variation of voltage e0 as a function of es has been plotted. From point 0 to point A on this graph, diode 3| is the only diode drawing current. From A to B, diodes 3| and 32 are both drawing current, from B to C diodes 3|, 32-, and 33 are drawing current, and from C to D diodes 3|, 32, 33 and 34 are all drawing current. As voltage es increases from zero to 40 volts, the voltage e0 is always a linear function of es up to the point where an additional diode starts drawing current. The current drawn through the additional diode is always of such magnitude and direction that the slope of the curve of Figure 2 is changed from positive to negative, or vice versa.

Thus far the discussion has been limited to what happens during that period in which the upper end of secondary winding I5 is positive with respect to the midtap of this winding. During the other half cycle of e5, when the upper end of I5 is negative with respect to the midpoint, diodes 3|, 32, 33, and 34 do not draw any current, but diodes 35, 36, 31, and 38 function in substantially the same manner as previously described for diodes 3|, 32, 33, and 34, and voltage e0 passes through substantially the same series of values that have been plotted in Figure 2 for negative values of es.

' Resistors 53 and 5| have resistance values much greater than that of resistor 49, so that the magnitude of the voltage developed across 49 is only slightly affected by any current flowing to ground through resistors 56 and 5|. To compensate -for these parallel currents it is necessary for 49 to have a resistance a little greater than 40,000 ohms, so that the resultant resistance from bus 4| to ground will be approximately 40,000 ohms. The ungrounded end of 49 is connected through resistor 50 to grid No. 3 (i. e. the third grid from the cathode) of vacuum tube El, and is also connected through resistor 5| to grid No. 3 of vacuum tube 62. Capacitors 65 and 66 are small radio-frequency by-pass capacitors, connected to ground.

The eiTects produced by the voltages applied to the No. 3 grids of tubes 6| and 62, and the effects produced by rectifiers ll and 12, hereinafter to be described, can best be understood after the functions of tubees 5|, 62, 63, and 64 are first explained. Tubes 6|, 62, 63, and 64 are of the type generally referred to as pentagrid converters. Grid No. 1 (i. e. the grid nearest the cathode) of 5|, and grid No 1 of 62 are connected to opposite ends of the tuned circuit composed of inductor 30 and condenser 8|. Grid No. 1 of 63 and grid No. 1 of 64 are connected to opposite ends of the tuned circuit composed of inductor 82, and condenser 83. Radio-frequency voltage from a suitable oscillator 84 is applied to the tuned circuit composed of 82 and 83 through a very small capacitor 85, which latter couples this tuned circuit relatively loosely to the oscillator.

The tuned circuit composed of and 8|, and the tuned circuit composed of 82 and 63 are both resonant at the frequency of oscillator 84. Coils 80 and 82 are located in such a manner that there is inductive coupling between them, as here shown by the arrow, and the degree of this coupling is such that the total voltage developed across 86 is equal to that developed across Furthermore, since the circuit composed of 80 and 8| is a resonant circuit, the total voltage consequently developed across 83 is approximately degrees out of phase with the voltage across 82. The phase of the voltage across coil 60 can be adjusted so as to be exactly 90 degrees from that across coil 62 by slightly varying the capacitance of condenser 8|.

Since the plates of tubes 6! and 62 nected in parallel to the unted circuit composed of condenser 86 and inductor 8?, tubes 6| and 62 are equivalent to, and capable of functioning as a balanced modulator. If grid No. 3 of tube 6| is at the same potential as' grid No. 3 of tube 52, no radio-frequency voltage will be produced across coil 87 and capacitor 86, because the radio-frequency voltage applied between grid No. 1 and cathode of tube 6| is degrees out of phase with the radio-frequency voltage applied between grid No. 1 and cathode of tube 62, and the gain of 6! is the same as that of 62, this action being that well known in the art of balanced modulators.

If the potential of grid No. 3 of tube 6| is varied in a positive direction from its normal potential, the output currents of tubes 3| and 62 will no longer balance out, because the gain of 6| will be greater than that of 62, and a radio-frequency voltage will appear across 87. If the potential of grid No. 3 of tube 5! is varied in a negative direction from its normal potential, a radio-frequency voltage will appear across coil 3?, but it will be 180degrees out of phase with the voltage which would have been produced if the potential of grid No. 3 direction from normal. The eiTect' of varying the potential of grid No. 3 of tube 62 is similar to that are (3011- had been varied in a positive s eep-721 of varying the potential of grid :No. .3 er tube 6!, except that, so far as the phase of ?the :output voltage .is concerned, :the effect .of raising tthe potential of grid :No. 3 of tube .52 is the 'same as that of lowering the potential of :gr'idNo. .3 of tubeffil.

In Figure .2 itwas indicated .thateo, the voltage across resistor ie, when plotted as 'aifunction of theitransforiner voltage 63, has a triangular .form. However this voltage is transmitted through the network composed of resistor '50, resistor I53, :and copper-oxide rectifiers ii :and :2, before being applied betweengrid and cathode ofztube 5.1. The copper-oxide rectifiers are of any suitable small instrument type having the characteristics Ehereinaiter described. A plurality, for example eleven, rectifiers of this type are vused'in seriesxto 'form rectifier unit 4 i, and eleven to form rectifier unit 32, and the two rectifiers .are so connected that the polarity of 'i! is opposite to that of 12. ItJis possible :to substitute a single resistor, having similar :IlOll-HYXQSJ. characteristics for these Irectifiers.

A D. C. bias voltage, of 2 volts, is applied between pointPz and ground, by means of bypassed resistor we, thus making point P2 positive relative'to ground. Therefore whenever the voltage-e across resistor ii) is 4 volts, point P1 is positive with respect to point P2 by an amount equal to 2 voltszminus .the vcltageaci'oss se. 0n thet'other' hand, whenithe voltage across 3-9 is zero,.point P1 isnegative with respect'to P2 by an amount equal to 2 volts minus'the voltage drop across resistor 50. Because of the non-linearity of the copperoxide rectifiers "ii and E2, the voltage-e1 between po nt P1 and point P2 varies as shownibylthe curve as: of Figure 3,130 which reference is now made, when '80 is varied. The curvature of this latter curve is caused by the fact that the resistance of the copper-oxide reetifiers decreases as the voltage across them increases.

Voltage e1 is applied between grid No. 3 and the cathode of tube 6i. At the same time, the voltage e2 developed across resistor 52 is applied between grid l-Zo. 3 and the cathode of tube 62,

and ez is directly proportional to voltage co, as indicated by curve 302 of :Figure 3.

Since the excitation voltages applied to the No, 1 grids of tubes El and 62 from coil 80 are 180 degrees out of phase with each other, the radio-frequency output voltage appearing across coil 81, due to the action of modulators 61 and 62, is proportional to the difference between (21 and-e2. This difference has been plotted as curve 303 of Figure 3. All of the curves shown in Figure 3 were plotted from data obtained in actual measurements of a physical embodiment of this invention. To obtain the data for the (er-e2) curve, a tube voltmeter was connected between points P1 and P3, voltage e0 was varied in small steps, and readings were taken of (er-e2), which is equivalent to the voltage between point P1 .and point P3. Resistor .50 was 130,000 ohms, 51 was M01100 ohms, 52 was 200,000 ohms, and 5.3 was 500,000 ohms. H and 12 each consisted of eleven Conant rectifiers, type BI-l, Series 160., connected in series. The circled points on the (@1-2) curve of Figure 3 represent measured values of (cl-e2), and also represent substantially the values that (cl-e2) would have if (e1-ez) were a cosine function of era, with maximum at-eo=+4 and minimum at en=0.

Reference is now made "to Figures 41:03, in-

clusive. It can be seen that (e1e2) is substantially a cosine function oieo, which :is the:condi-- itioni'desired, .a's later become :appareut. zFigure shows (er- 22) as .a Ifimction not the titransformer voltage es. Since-emisza linearfimction "of ice, except Where u'eversals occur :in the slope of tthe curve shown in Figure '2, :and rev-e2) is substantially ta cosine ;function of en, ithen (e1ez) is substantially a cosine efunction 'of 'esgasindicated in Figure 4.

Diod'esiSl Tto :98 inclusive, together =withvo1tage dividers 1'03 and 104, ifunction in .the'same :man- :ner as do diodes 3| to -38 inc1usive, except athat diodes 9] to i98are biased diiferently. Each tap :*on voltage divider 103 is located at .a :point :5 volts higher in potential than-the corresponding tapon divider I 8,:and eachtap on voltage divider 11,4518 located'at a point 5ivolts lower in potential than 'the'correspondin point on divider [9. :Because of this arrangement of biases, relative to one another, the voltage across resistor l -M'varies in the manner shown in :Figure 5, instead of the manner :shown in Figure '2. The curve :of Figure 5 is substantially the same as thatshown in Figure v2, except "that it has been shitted to thefleft, 580 that when is zero,lthe voltagezacross resistor H4 is z volts, instead of zero. 6 shows (e3'e4) as :a .function of 3s. Evidently, (63-84) is a sine'function of "85, instead of being a cosine function, as'is :(erez) (Fig. 4) This causes :a .phase displacement between the two transmission channels.

:Now considering the conditions existing in Tithe common plate circuit of tubes 61 and 62 attire instant when as is zero, reference to Fig. ishows that (en-e2) issat a negative peak, so that the resultant radio-frequency rcurrent, "produced wholly by tubes 5| and .62 in the :coil 81 maybe represented by vector 11, in Fig."7. .At the same instant (33-89 is zero,:so tubes :63 and 54 are not producing any radio-frequency .plat'e current. Therefore a'vector I2, representing the current .due to these last :tubes, :is at this instant, 'zero. The ii'esultant radio-frequency iplate current is therefore equal in magnitude and phase :to 11-. At the instant when e5=+2;5 volts the -output currentlr of tubes iil ian'd62 is equal (in amplitude to the output current 112 of-tubes 63 and 64, because '(e1- ez)=-(ea-e4)., as' a comparison :of Fig. 4 and Fig. '6 will-show. The resultant RAF. plate :current IT -th'eref ore, as shown in 8. When voltage es arrives :at a value :of '5 volts, (er-e2) =0, and (es-e4) is :a maximum. The resultant current .is therefore equal "to 12, which has the phase and value indicated in Fig. 9, while I1, is IIOWLZQIO. As voltage 'escon'tinues to increase,-the resultant current .=IT'is continuously shifted in phase, .in .a clockwise dilectiDIl, mitil finally, .vvhen es becomes +35 volts-"a total phase shift of 1% cyclesy-or 630 degrees, has ibeen :produced.

It is 'obvioustha't in order to makethe resultant vector IT rotate through 360 degrees or more without any change in'length,:itis necessary that each of the component vectors I1, and ilz s'hall in turn undergo a reversal of direction, corresponding to aphase reversal, when the .resultarrt passes from each quadrant into the next succeeding quadrant, during the course of the modulation cycle. These reversals are able to occurbecause of the use of two balanced modulators, each :of whose *RrF. -output currents can ibe reversed in phase byrevers'ingithe :polarity sofvolta'ge ('61-- ea) or voltage (er-er).

Thirty five volts is the maximum valuejwhich the voltage 657C311]. have without scausing distortion in the transmitted signal, in theparticulur embodiment here shown. When es returns from +35 volts to zero, the phase of the resultant current IT varies in a counterclockwise direction instead of a clockwise direction. During that part of the cycle in which es is negative, the operation is similar to that for the positive half cycle. In either case, varying es in a positive direction always causes the phase of IT to vary in a clockwise direction, and vice versa.

Coil 8-3 can of course be connected to an amplifier or a frequency multiplierand the radiofrequency signal voltage can be amplified, or the frequency can be multiplied, in substantially the same manner as in other frequency-modulation transmitters.

As previously mentioned, when the amplitudephase-frequency system of modulation is used in broadcast transmitters, the phase deviation produced by the modulation system is usually limited to about :11.5 degrees, or 10.2 radian, to avoid any serious distortion. In my system, the maximum phase deviation that can be produced by the embodiment here shown, without serious distortion, is $630 degrees, or :11 radians, which is about 54.8 times as great as the figure of 11.5 degrees given above. It is obvious that several stages of frequency multiplication can be eliminated by the use of my system. Furthermore the large phase deviation obtained in my modulation system makes it unnecessary to use heterdoyning or dual channel operation. For example, let it be supposed that the lowest audiofrequency to be transmitted is 20 cycles/sec. The frequency deviation produced by an ll-radian phase deviation at this modulating frequency would be 20x11, or 220 cycles/sec. If the average output frequency of the transmitter is to be 100,000 kc. and the frequency of the oscillator is 200 kc., a total frequency multiplication of 500:1 is needed. This frequency multiplication will increase the frequency deviation to a value of 220x500, or 110,000 cycles/sec, which is more than enough to meet the present requirements in the art, as fully described in an earlier portion of this specification.

The method of selecting resistance values for resistors 2| to 28 is now explained. The values of 2| and 49 are arbitrarily selected as 60,000 ohms and 40,000 ohms respectively, thus causing the voltage e across 49 to be 4 volts when es is volts. It has already been decided that when es has risen to volts, the current through 49 shall drop to zero, so that en will be zero (see Fig. 2). At this point the voltage across resistor 2| will be 20 volts, and the current through 2| will be 20/60,000=0.333 milliampere. Diode 32 starts drawing current when es is +10 volts, and by the time es is +20 volts the voltage across resistor 22 is 20 14=6 volts. Since the current drawn through 22 must at this time be equal to that through 2 I, in order to obtain zero resultant current through 49, the resistance of 22 must be 6/(0.333 10- =18,000 ohms.

As indicated in Fig. 2, the network is to be designed so that when es is volts the voltage e0 will be 4 volts. When es rises above +20 volts, diode 33 starts drawing current, and when es is +30 volts the voltage across 23 is 30-20-4=6 volts. At the same time, the current through 2| is (30-4)/60,000=0.4333 milliamperes, and the current through 22 is (3014+4) /18,000=1.1111 milliamperes. To obtain a voltage of 4 volts across resistor 49 we must have a resultant cure n Ur) 10 rent of 0.1000 milliampere through 49. To obtain this resultant current, the current through 23 must be 0.1000-0.4333+1.111=0.7778 milliampere. Since the voltage across 23 is 6 volts, the resistance of 23 must be 6/ (0.7778 10 =7,718 ohms.

When es rises above +30 volts, diode 34 starts drawing current, and by the time 25 has reached a value of +40 volts the voltage across 24 is (40-34) :6 volts. At the same time the current through 2! is 40/60,000=0.6667 milliampere, the current through 22 is (40 l4) /18,000=1.4444 milliamperes, and the current through 23 is (40-20) /7,718 which equals 2.5913 milliamperes. The resultant current through 49 must be zero, and therefore, the current through 24 must be 0.6667+1.444+Z.59l3=1.8136 milliamperes. To obtain this current it is necessary that 24 must have a resistance of 6/ (1.8136 X 10 =3,308 ohms.

The calculations made so far have pertained to diodes 3|, 3'2, 93, and 34, which draw current during the half cycle when the upper end of secondary winding I5 is positive with respect to the midpoint. During the other half cycle the operation is similar to that on which the above calculations were based, but diodes 35, 39, 31, and 38 draw current instead of the above mentioned diodes. Therefore resistor 25 should be equal in value to 2 l, as should be equal to 22, 27 should be equal to 23, and 28 should be equal to 24. The resistors in series with diodes 9! to 98 inclusive should be equal in value to the corresponding resistors used in series with diodes 3! to 38 inclusive.

In determining the values of resistance as above calculated, certain factors. which slightly affect the result are disregarded for the sake of simplicity. One of these is the contact potential of the diodes. This contact potential, which is usually only a fraction of one volt, produces the same effect as though a battery of this voltage were connected in series with the diode. Other factors which affect the result are the respective voltage drops produced by the diode currents in flowing through voltage dividers l8, I9, I03 and I04, although it is intended that the resistors used in the voltage dividers shall have a very much lower resistance than resistors 21 to 28 inclusive. To allow for these factors and any other factors affecting the results, it is best to make all of the series resistors adjustable, and all taps on the voltage dividers adjustable. After all resistors and taps have been adjusted according to the above calculated values, an alternating Voltage should be applied to the input of amplifier I2, and adjusted in magnitude so as to produce a voltage of 80 volts peak across each entire secondary winding M or 15. The voltage across these windings should be applied to the horizontal deflection plates of a cathode-ray oscilloscope, or other equivalent device, using amplification if necessary. The voltage e0 across resistor 49 should be amplified, and applied to the vertical deflection plates of the same oscilloscope. The pattern on the screen of the oscilloscope should then appear the same as shown in Fig. 2. If adjacent peaks are not all separated by the same horizontal distance, or are not of equal height, the necessary corrections can be made by adjusting the taps on voltage dividers l8 and 19, or by adjusting the resistors in series with the various diodes.

Because of the fact that the various diodes do not all draw current during the entire half of? clarity cycle; the load on the secondary windings of transformer i3 is not a linear load. This tends to prevent. voltage 6s. from having exactly thesame wave form as the input voltagerof amplifier i2. This effect can: be greatly reduced by loading transformer is with linear resistors 21-5 and 2H5; Any remaining efiects produced by the non-linear load can be compensated by adjusting the taps on the. voltage-dividers,.and the resistors in series with the diodes.

The calculated values of' resistance for the resistors in series with the diodes, and the calculated position. ofthe taps on the voltage dividers, should be considered as first approximations only, since changes may have to be made when the above described test is performed, and the final adjustments may be considerably different from those indicated. by the calculations.

Obviously, it would be possible to use copperoxide, selenium, or other similar rectifiers. instead of the diodes here shown, although theresistance values of. the. various resistors used. in. various parts of the associated network might have to. be somewhat different, because of. the fact that therectifiers. for example the copperoxide or selenium. type,. do not have exactly the. same characteristics as do thermionic diodes, aswelllknownin. the art. Such rectifiers need not be strictly linear, as variations are substantially nullified by the high resistances connected in series therewith.

Tubes 6 l 52, 63,,and 64, havebeen represented in Fig. l as pentagrid converter tubes. For ex.- ample, tubes of the well known type GSA? or of the type GBES could be used. Instead of using pentagrid. converter tubes it would be possible to use various other. types of tube, for example the type GSJ'I' sharp-cutoii pentod'e. In using this last type of tube applied to the control grid (i;. e., the grid nearest the cathode), while the oscillator voltage canbe applied. to the. screen grid thereof.

Elements not specifically'described in. the lower transmission. channel, correspond to those shown in the upper channel, and detailed description thereof is therefore not needed. The. various grounds; bias resistors, by-pass condensers, etc, are likewise not discussed in detail, for purposes of clarity; as their respective functions will be apparent to one skilled in the art.

Reference is now made toFig. 10,- a schematic showing of a portion of an alternative system which makes it possible to produce an even greater phase deviation than that produced by the circuits of Fig; 1'. The portions not shown in Fig. lO-may be identical with those illustrated in Fig. l. Resistors 4'9 and H4- correspond'to the resistors having thesame reference numerals in'Fig. 1 The voltages-producedacross these resistors have the same form as' the voltages across the corresponding resistors in Fig. 1', i; a. they: are as. shown in: Fig; 2' and Fig; 5-. The circuits and: elements used for. producing these voltages across. resistors 49' and H4 may be identicaliwith; those shown. in Fig. 1, and operatein the same manner; as already described in connection: with Figs; 1, 2,,and. 5. For purposesthey are omitted iiLFig; 10.

The voltage; across resistor 49 is amplified by triode EM, which is a. linear amplifier, and is reproduced in; amplified form. across resistor 303; Some of; the voltage. across resistor 3&3: is. applied, through; the voltagedivider consisting of:

the modulating voltage can beresistors 3M; and 3.05;,to the; grid; circuit; or title ode 302.. This voltage is reproduced in. ampii fled. form,. across resistor 3.96;. but the. voltage: across resistor 365- is.\ mil-degrees out of phase: with the, voltage across resistor 353:, because of the phaserinverter action oftriode 352:. The: resistance values of: resistors 30.4: and 305. are chosen. so that. the. voltage. produced across: re;' sistor 30B is equal in magnitude to that produced across resistor 303;. While triodes 38! and? 362 are shown. as located; within asingle envelope, it. is to be understood that discrete tubes. may be" employed:

The: voltage of bias.- battery 3M.- and the'values of the various resistors are also chosen so that.

when. the voltage across resistor 49.- is halfway between zeroand maximunnpoint ii-Mwill beatthe. same potential as point 3M. Since. these values depend, inter alia, upon: tube character istics,. they must be ascertained, in anygiven. case by. methods well known; in the art. The- D..-C.. potential of. point 309 is. adjusted in a. similar manner, so that. it is equal to the:poten-- tial of points 3L0. and. sit. at the instant when. the. potential. of point. 3.1.0 is equal. to the: potential of point 3|.L. Resistor. 3.4.9. forms part of: the. network used. to. secure the proper voltage relationships... This. enables rectifiers 301. and 3.08 to function as a full-wave. rectifier,. so. thatv the voltage. across output resistor 3&2: has theform shown in Fig..12.. Fig. 1]., which. has been plotted directly above. 12. for purposes. of comparison, shows the. voltage across input re.-- sistor. 9. The number of; peaks in Fig. 12. is-

; twice as great as the. number. in Fig. 11.. Rectie fiers 307 and 308 are. preferably, though. not. necessarily, of the germanium crystal typasuch as the type D134.

The network composed of resistors 55,. 52, 53,.

- and 54;. and" rectifiers H and 12, functions. in

the: same manner. as the similar network in Fig. 1, producing voltages having the. same form as. voltages e1 and c2 of Fig; 1. These. voltages are applied to the N0. 3' grids. of balanced modulators- 6i and 62, which likewise function in the same manner as the corresponding modulators of Fig. I.

Theaction of triodes 32f. and'322, rectifiers 3211 and 328, and modulator tubes 63. and. Gi l, is. the same as that. of' the corresponding elements in the-dipper channel, i: e., thasame as. 353i, 302', 30], 3%, 6t, andGZ; respectively. The rectifier net.- work between resistor: 3'22and. tubes 63 and 64 maybe substantially identical in structure and performancewith that" used in the upper channel'-, and-"therefore i's-not described in. detail; Fig. li3' sliowsthe voltage across resistor322. It can be seen that the voltage across resisitor 322' is displaced by aquarter cyclefrom the voltage acrossresistor 3E2; B'ecauseof the frequencydoubling action produced by rectifiers 321 and 328- it is -necessary that'the-volt'age across resistor I-I'Aibedisplaced from that across resistor iiiby only one-eighth cycle; instead of one-quarter cycle. This may be" accomplishedby adjusting the circuitelements so as toisecure certain-biases differentfrom those indicated in Fig. 1'. Specificallyaomvolt'age divider 1-03 the points in Fig; 1 marked-15 n, +5-v., -i--l9" vi, and +39"v., should now'havepotential's of -17.5 v., +2.5 v., +l6.5 v., and.+36:5iv.,.respective1y; Similar-1y, on voltage divider IIM thepoints in Fig. 1- marked 25v., 5.v-., +-9-v., and +29-v.-, should now have-potentialsof -22i5v., --2.5 v-., +Il-L5 v., and: +1 .5 prospectively:

Since the number of peaks in Fig 12 and Fig. 13 is twice as great as the number in Figs. 2 and 5, it is evident that for an audio-frequency voltage es having a peak value of 35 volts, the total phase deviation in the radio-frequency current in coil L-3 will be twice as great, when the system shown in Fig. 1G is used, as it is when the system shown in Fig. 1 is used.

It will be obvious to one skilled in the art that by adding more amplifiers similar to 39 5, SM, 32 l and 322, and more rectifiers similar to Bill, 388, 321 and 328, together with their associated resistors, it is possible to introduce additional rectifications which, in cascade, will result in still further phase deviation of the transmitted cu.r-'

rent.

From the foregoing description it can be seen that the essential points involved in this invention may be summarized as follows.

This invention has the advantage of providing a frequency-modulated transmitter of the modulation type in which the amount of frequency multiplication is greatly reduced from that found necessary in the prior art. Additionally, the necessity of frequency reduction by heterodyning is obviated,

The advantages just enumerated are secured by the employment of devices which allow the initial phase deviation to extend over a substantially unlimited range, instead of confining it to a maximum of about X; radian, as has been done in the prior art. This increased range is secured without transcending the degree of distortion considered allowable.

In the system of this invention the audio-frequency modulating signals are modified so as to act upon the two radio-frequency circuits carrying currents 90 out ofphase with one another, in two distinctive manners, whereby the distortion which occurs when a balanced modulator of the conventional type is employed is substantially eliminated, even though the phase shift may reach a value of many radians.

Expressed in other terms, this system simultaneously modulates the two vectors involved in modulation, in such fashion that distortion is substantially absent. Furthermore, when the resultant of these two vectors leaves the first quadrant, the modulation undergoes a corresponding alteration. -This same action takes place as'the resultant of the vectors enters each succeeding quadrant, so that there is no theoretical limit to the number of times that the resultant of the vectors can describe a complete 360 revolution.

While the two balanced modulators are here shown as excited 90 out-of-phase with one another, by high frequency oscillator 84, it will be evident that both modulators could be excited in phase with each other, provided that the respective outputs of the modulators are shifted relatively to one another, so as to cause the desired 90 out-of-phase relationship, before these two outputs are combined with each other. The equivalence of these alternative modes of connec tion is well known in the art.

While there have been shown and described certain embodiments of this invention it is to be understood that many variations thereof will be apparent to those skilled in the art and accordingly the scope of this invention is limited only by the hereunto appended claims.

What is claimed as new and desired to be secured by Letters Patent of the United States, is:

1. In combination, an oscillator, a first balanced modulator receiving voltage from said oscillator, a second balancedmodulator also receiving voltage from said oscillator, means for causing said last voltage to be ninety degrees out of phase with the first mentioned voltage, a unitary source of audio-frequency signalling voltage, means for reversin the phase of the output current of said first balanced modulator in accordance with a function of the magnitude of said signaling voltage at a plurality of differing values, means for subsequently reversing the phase of the output current of said second balanced modulator inaccordan-ce with a similar function, means for again reversing the phases of the output currents of the balanced modulators but in opposite sequence to the previous reversals, and. means for combining the output currents of said balanced modulators.

2. In combination, an oscillator, a first balanced modulator receiving voltage from said oscillator, asecond balanced modulator receiving voltage from said oscillator, means for causing said last voltage to be ninety degrees out of phase with the first mentioned voltage, a unitary source of audio-frequency signalling voltage, meansfor producing a series of reversals in the phase of the output currents of said balanced modulators in accordance with a function of the magnitude of said audio-frequency signaling voltage when said voltage is altering in value upon a single side only of the zero line, said reversals occurring in a predetermined sequence with the reversals of said first balanced modulator occurring at different times from the reversals of said second balanced modulator, means for reversing the sequence in which said reversals occur, and means for combining the output currents of said balanced modulators.

3. In combination, an oscillator, a first balanced modulator receiving voltage from said oscillator, a second balanced modulator receiving voltage from said oscillator, means for causing said last voltage to be ninety degrees out-ofphase with the first mentioned voltage, a source of signal voltage, means for causing the amplitude of the output current of said first balanced modulator to vary in proportion to a sine function of the instantaneous magnitude of said signal voltage, means for causing the amplitude of the output current of said second balanced modulator to vary in proportion to another sine function of the instantaneous magnitude of said signal voltage, said other sine function being displaced in phase from said first mentioned sine function, and-means for combining the output currents of said balanced modulators.

4. A frequency modulated transmitter, including means for producing audio-frequency voltage, means for separating said voltage into two channels, means in the first channel for transforming said voltage into a first wave form triangular with respect to the instantaneous magnitude of said audio-frequency voltage, means in the second channel for transforming said voltage into a second triangular wave form, displaced in phase relationship with respect to said first triangular wave form, means in each channel for transforming the respective triangular wave forms therein into wave forms sinusoidal with respect to the instantaneous magnitude of said audio-frequency voltage, two balanced modulators each modulated by the output of one of said channels, means for feeding carrier voltage to both said modulators, means for withdrawing the respective outputs of both said modulators, means for rendering one output substantially 90 out-of phase. with: respect to: therother out?- put, and means for. combining; both said-outi-of-- phase outputs.

5; Atransmitter according to claimA, inwhich said means. for transforming ages into triangular wave form; comprises a plurality of rectifiers, means for: biasing each rectig r to a different voltagasaid rectifiersbeing o positely disposed,

triangular waveforms. percycle of the audiofrequency voltage is a. function of the amplitude of the audio-frequency voltage feeding said rect-ifiers.

6. A transmitter accordingto claimA, in which said means for. transforming. said triangular Wave forms into sinusoidal: forms comprises a plurality ofrectifiers having nonelinear characteristics;- connected in oppositely: poled: pairs.

'1'. A transmitter according. to claim. 4, addiltionall'y including means in each channel for doubling the number of cycles of said triangular last means comprising wave form voltage, said phase inversion means, two rectifiers, each located' in one output" circuit-of said: phase. inversion means, so as to be 180 out. of phase with one another, and means for biasingrectifiers; whereby they: function alternately.

8-. The method of frequency modulationrwhicn includes the stepsofproducing two carrier currents 90 outofphase with each other, producing an audio-frequency voltage; transforming said audio-frequency voltage into two: voltages,

the respective magnitudes of which are linear functions of" saidaudio-frequency voltage, the respective slopesof said: linear functions: re-

versing in algebraic sign at a. given number of difiering predetermined voltage magnitudes of said audio-frequency voltage, transforming said two voltages-intorespective sine functions which have peaks occurring. at different moments; modulating each of said carrier currents bya different one of said two voltages, and combiningsaid two modulated carrier-currents.

9: The method of' frequency modulation according to claim 8", including the additional step of doubling thenumber of reversals-of said linear functions with respect/to the number of voltage magnitudes, whereby the phase deviation secured in said combined" modulated carrier currents is likewise doubled.

the: respective volt-- and being connected tov a cqinmonloadwhereby the: number of complete said last.

-said voltages at predetermined magnitudes. of

said audio-frequency voltage, said predetermined; magnitudes being different for each of saidE two voltages; transforming said two voltages into-re:-- spective sine functions, modulating each ofsaid: carrier currents by a difierent one of the two transformed voltages, and combining the two.

modulated carrier currents.

12. A combination: according to claim 1, in; which saidmeans for reversing the phase of the:

D output of said first balanced modulator include a:

plurality of difierently biased rectifiers and a: plurality of non-linear resistances.

13. A combination'according to claim Z-wherein said means for producing a series of reversals; in the phase of the output current of. at least one: of said balanced modulators include an 81861311?- cal resistance and rectifier network yielding, an; output voltage which is a periodic function oi the magnitude of the input voltage thereof,

, means for changing said output voltageto. a sine function of said input voltage, and means for applying said changed output voltage to one of said balanced modulators.

FREDERICK W. FRINK.

REFERENCES CITED The following references are of record in the file of this patent:

UNITED STATES PATENTS Number 7 Name Date 1,719,052 Green July 2, 1929' 1,747,160 Carpe Feb. 18, 1930 2,289,564 Wrathall July 14', 1942 2,424,971 Davey- Aug. 5, 194'? 

